Tunable compensation circuit for filter circuitry using acoustic resonators

ABSTRACT

Tunable filter circuitry includes a series acoustic resonator between first and second nodes and a compensation circuit in parallel with the series acoustic resonator. The compensation circuit includes first and second inductors coupled in series between the first node and the second node, wherein the first inductor and the second inductor are negatively coupled with one another and a common node is provided between the first and second inductors. The compensation circuit also includes first and second shunt acoustic resonators, which are coupled in parallel with one another between the common node and a fixed voltage node. A first variable capacitor is also coupled between the common node and the fixed voltage node, wherein changing a capacitance of the first variable capacitor changes a bandwidth of a passband of the filter circuitry.

RELATED APPLICATIONS

This application claims the benefit of provisional patent applicationSer. No. 62/317,675, filed Apr. 4, 2016, the disclosure of which isincorporated herein by reference in its entirety.

This application is a Continuation-in-Part of U.S. utility patentapplication Ser. No. 15/275,957, filed Sep. 26, 2016, which claims thebenefit of provisional patent application Ser. No. 62/232,746, filedSep. 25, 2015; the disclosures of which are incorporated herein byreference in their entireties.

FIELD OF THE INVENTION

The present disclosure relates to acoustic resonators and in particularto a tunable compensation circuit for filter circuitry using acousticresonators.

BACKGROUND

Acoustic resonators, such as Surface Acoustic Wave (SAW) resonators andBulk Acoustic Wave (BAW) resonators, are used in many high-frequencycommunication applications. In particular, SAW resonators are oftenemployed in filter networks that operate frequencies up to 1.8 GHz, andBAW resonators are often employed in filter networks that operate atfrequencies above 1.5 GHz. Such filters need to have flat passbands,have steep filter skirts and squared shoulders at the upper and lowerends of the passband, and provide excellent rejection outside of thepassband. SAW- and BAW-based filters also have relatively low insertionloss, tend to decrease in size as the frequency of operation increases,and are relatively stable over wide temperature ranges. As such, SAW-and BAW-based filters are the filter of choice for many 3rd Generation(3G) and 4th Generation (4G) wireless devices and are destined todominate filter applications for 5th Generation (5G) wireless devices.Most of these wireless devices support cellular, wireless fidelity(Wi-Fi), Bluetooth, and/or near field communications on the samewireless device and, as such, pose extremely challenging filteringdemands. While these demands keep raising the complexity of wirelessdevices, there is a constant need to improve the performance of acousticresonators and filters that are based thereon.

To better understand acoustic resonators and various terminologyassociated therewith, the following provides an overview of a BAWresonator. However, the concepts described herein may employ any type ofacoustic resonator and are not limited to SAW- and BAW-based resonators.An exemplary BAW resonator 10 is illustrated in FIG. 1. The BAWresonator 10 generally includes a substrate 12, a reflector 14 mountedover the substrate 12, and a transducer 16 mounted over the reflector14. The transducer 16 rests on the reflector 14 and includes apiezoelectric layer 18, which is sandwiched between a top electrode 20and a bottom electrode 22. The top and bottom electrodes 20 and 22 maybe formed of Tungsten (W), Molybdenum (Mo), Platinum (Pt), or likematerial, and the piezoelectric layer 18 may be formed of AluminumNitride (AlN), Zinc Oxide (ZnO), or other appropriate piezoelectricmaterial. Although shown in FIG. 1 as each including a single layer, thepiezoelectric layer 18, the top electrode 20, and/or the bottomelectrode 22 may include multiple layers of the same material, multiplelayers in which at least two layers are different materials, or multiplelayers in which each layer is a different material.

The BAW resonator 10 is divided into an active region 24 and an outsideregion 26. The active region 24 generally corresponds to the section ofthe BAW resonator 10 where the top and bottom electrodes 20 and 22overlap and also includes the layers below the overlapping top andbottom electrodes 20 and 22. The outside region 26 corresponds to thesection of the BAW resonator 10 that surrounds the active region 24.

For the BAW resonator 10, applying electrical signals across the topelectrode 20 and the bottom electrode 22 excites acoustic waves in thepiezoelectric layer 18. These acoustic waves primarily propagatevertically. A primary goal in BAW resonator design is to confine thesevertically propagating acoustic waves in the transducer 16. Acousticwaves traveling upward are reflected back into the transducer 16 by theair-metal boundary at the top surface of the top electrode 20. Acousticwaves traveling downward are reflected back into the transducer 16 bythe reflector 14 or by an air cavity, which is provided just below thetransducer in a Film BAW Resonator (FBAR).

The reflector 14 is typically formed by a stack of reflector layers (RL)28, which alternate in material composition to produce a significantreflection coefficient at the junction of adjacent reflector layers 28.Typically, the reflector layers 28 alternate between materials havinghigh and low acoustic impedances, such as tungsten (W) and silicondioxide (SiO₂). While only five reflector layers 28 are illustrated inFIG. 1, the number of reflector layers 28 and the structure of thereflector 14 varies from one design to another.

The magnitude (Z) and phase (φ) of the electrical impedance as afunction of the frequency for a relatively ideal BAW resonator 10 isprovided in FIG. 2. The magnitude (Z) of the electrical impedance isillustrated by the solid line, whereas the phase (φ) of the electricalimpedance is illustrated by the dashed line. A unique feature of the BAWresonator 10 is that it has both a resonance frequency and ananti-resonance frequency. The resonance frequency is typically referredto as the series resonance frequency (f_(s)), the anti-resonancefrequency is typically referred to as the parallel resonance frequency(f_(p)). The series resonance frequency (f_(s)) occurs when themagnitude of the impedance, or reactance, of the BAW resonator 10approaches zero. The parallel resonance frequency (f_(p)) occurs whenthe magnitude of the impedance, or reactance, of the BAW resonator 10peaks at a significantly high level. In general, the series resonancefrequency (f_(s)) is a function of the thickness of the piezoelectriclayer 18 and the mass of the bottom and top electrodes 20 and 22.

For the phase, the BAW resonator 10 acts like an inductance thatprovides a 90° phase shift between the series resonance frequency(f_(s)) and the parallel resonance frequency (f_(p)). In contrast, theBAW resonator 10 acts like a capacitance that provides a −90° phaseshift below the series resonance frequency (f_(s)) and above theparallel resonance frequency (f_(p)). The BAW resonator 10 presents avery low, near zero, resistance at the series resonance frequency(f_(s)) and a very high resistance at the parallel resonance frequency(f_(p)). The electrical nature of the BAW resonator 10 lends itself tothe realization of a very high Q (quality factor) inductance over arelatively short range of frequencies, which has proved to be verybeneficial in high-frequency filter networks, especially those operatingat frequencies around 1.8 GHz and above.

Unfortunately, the phase (φ) curve of FIG. 2 is representative of anideal phase curve. In reality, approaching this ideal is challenging. Atypical phase curve for the BAW resonator 10 of FIG. 1 is illustrated inFIG. 3A. Instead of being a smooth curve, the phase curve of FIG. 3Aincludes ripple below the series resonance frequency (f_(s)), betweenthe series resonance frequency (f_(s)) and the parallel resonancefrequency (f_(p)), and above the parallel resonance frequency (f_(p)).The ripple is the result of spurious modes, which are caused by spuriousresonances that occur in corresponding frequencies. While the vastmajority of the acoustic waves in the BAW resonator 10 propagatevertically, various boundary conditions about the transducer 16 resultin the propagation of lateral (horizontal) acoustic waves, which arereferred to as lateral standing waves. The presence of these lateralstanding waves reduces the potential Q associated with the BAW resonator10.

As illustrated in FIG. 4, a border (BO) ring 30 is formed on or withinthe top electrode 20 to suppress certain of the spurious modes. Thespurious modes that are suppressed by the BO ring 30 are those above theseries resonance frequency (f_(s)), as highlighted by circles A and B inthe phase curve of FIG. 3B. Circle A shows a suppression of the ripple,and thus of the spurious mode, in the passband of the phase curve, whichresides between the series resonance frequency (f_(s)) and the parallelresonance frequency (f_(p)). Circle B shows suppression of the ripple,and thus of the spurious modes, above the parallel resonance frequency(f_(p)). Notably, the spurious mode in the upper shoulder of thepassband, which is just below the parallel resonance frequency f_(p),and the spurious modes above the passband are suppressed, as evidencedby the smooth or substantially ripple free phase curve between theseries resonance frequency (f_(s)) and the parallel resonance frequency(f_(p)) and above the parallel resonance frequency (f_(p)).

The BO ring 30 corresponds to a mass loading of the portion of the topelectrode 20 that extends about the periphery of the active region 24.The BO ring 30 may correspond to a thickened portion of the topelectrode 20 or the application of additional layers of an appropriatematerial over the top electrode 20. The portion of the BAW resonator 10that includes and resides below the BO ring 30 is referred to as a BOregion 32. Accordingly, the BO region 32 corresponds to an outer,perimeter portion of the active region 24 and resides inside of theactive region 24.

While the BO ring 30 is effective at suppressing spurious modes abovethe series resonance frequency (f_(s)), the BO ring 30 has little or noimpact on those spurious modes below the series resonance frequency(f_(s)), as shown by the ripples in the phase curve below the seriesresonance frequency (f_(s)) in FIG. 3B. A technique referred to asapodization is often used to suppress the spurious modes that fall belowthe series resonance frequency (f_(s)).

Apodization tries to avoid, or at least significantly reduce, anylateral symmetry in the BAW resonator 10, or at least in the transducer16 thereof. The lateral symmetry corresponds to the footprint of thetransducer 16, and avoiding the lateral symmetry corresponds to avoidingsymmetry associated with the sides of the footprint. For example, onemay choose a footprint that corresponds to a pentagon instead of asquare or rectangle. Avoiding symmetry helps reduce the presence oflateral standing waves in the transducer 16. Circle C of FIG. 3Cillustrates the effect of apodization in which the spurious modes belowthe series resonance frequency (f_(s)) are suppressed, as evidence bythe smooth or substantially ripple free phase curve below the seriesresonance frequency (f_(s)). Assuming no BO ring 30 is provided, one canreadily see in FIG. 3C that apodization fails to suppress those spuriousmodes above the series resonance frequency (f_(s)). As such, the typicalBAW resonator 10 employs both apodization and the BO ring 30.

As noted previously, BAW resonators 10 are often used in filter networksthat operate at high frequencies and require high Q values. A basicladder network 40 is illustrated in FIG. 5A. The ladder network 40includes two series resonators B_(SER) and two shunt resonators B_(SH),which are arranged in a traditional ladder configuration. Typically, theseries resonators B_(SER) have the same or similar first frequencyresponse, and the shunt resonators B_(SH) have the same or similarsecond frequency response, which is different from the first frequencyresponse, as shown in FIG. 5B. In many applications, the shuntresonators B_(SH) are detuned versions of the series resonators B_(SER).As a result, the frequency responses for the series resonators B_(SER)and the shunt resonators B_(SH) are generally very similar, yet shiftedrelative to one another such that the parallel resonance frequency(f_(p,SH)) of the shunt resonators approximates the series resonancefrequency (f_(s,SER)) of the series resonators B_(SER). Note that theseries resonance frequency (f_(s,SH)) of the shunt resonators B_(SH) isless than the series resonance frequency (f_(s,SER)) of the seriesresonators B_(SER). The parallel resonance frequency (f_(p,SH)) of theshunt resonators B_(SH) is less than the parallel resonance frequency(f_(p,SER)) of the series resonators B_(SER).

FIG. 5C is associated with FIG. 5B and illustrates the response of theladder network 40. The series resonance frequency (f_(s,SH)) of theshunt resonators B_(SH) corresponds to the low side of the passband'sskirt (phase 2), and the parallel resonance frequency (f_(p,SER)) of theseries resonators B_(SER) corresponds to the high side of the passband'sskirt (phase 4). The substantially aligned series resonance frequency(f_(s,SER)) of the series resonators B_(SER) and the parallel resonancefrequency (f_(p,SH)) of the shunt resonators B_(SH) fall within thepassband. FIGS. 6A through 6E provide circuit equivalents for the fivephases of the response of the ladder network 40. During the first phase(phase 1, FIGS. 5C, 6A), the ladder network 40 functions to attenuatethe input signal. As the series resonance frequency (f_(s,SH)) of theshunt resonators B_(SH) is approached, the impedance of the shuntresonators B_(SH) drops precipitously such that the shunt resonatorsB_(SH) essentially provide a short to ground at the series resonancefrequency (f_(s,SH)) of the shunt resonators (phase 2, FIGS. 5C, 6B). Atthe series resonance frequency (f_(s,SH)) of the shunt resonators B_(SH)(phase 2), the input signal is essentially blocked from the output ofthe ladder network 40.

Between the series resonance frequency (f_(s,SH)) of the shuntresonators B_(SH) and the parallel resonance frequency (f_(p,SER)) ofthe series resonators B_(SER), which corresponds to the passband, theinput signal is passed to the output with relatively little or noattenuation (phase 3, FIGS. 5C, 6C). Within the passband, the seriesresonators B_(SER) present relatively low impedance, whereas the shuntresonators B_(SH) present a relatively high impedance, wherein thecombination of the two leads to a flat passband with steep low- andhigh-side skirts. As the parallel resonance frequency (f_(p,SER)) of theseries resonators B_(SER) is approached, the impedance of the seriesresonators B_(SER) becomes very high, such that the series resonatorsB_(SER) essentially present themselves as open at the parallel resonancefrequency (f_(p,SER)) of the series resonators (phase 4, FIGS. 5C, 6D).At the parallel resonance frequency (f_(p,SER)) of the series resonatorsB_(SER) (phase 4), the input signal is again essentially blocked fromthe output of the ladder network 40. During the final phase (phase 5,FIGS. 5C, 6E), the ladder network 40 functions to attenuate the inputsignal, in a similar fashion to that provided in phase 1. As theparallel resonance frequency (f_(p,SER)) of the series resonatorsB_(SER) is passed, the impedance of the series resonators B_(SER)decreases and the impedance of the shunt resonators B_(SH) normalizes.Thus, the ladder network 40 functions to provide a high Q passbandbetween the series resonance frequency (f_(s,SH)) of the shuntresonators B_(SH) and the parallel resonance frequency (f_(p,SER)) ofthe series resonators B_(SER). The ladder network 40 provides extremelyhigh attenuation at both the series resonance frequency (f_(s,SH)) ofthe shunt resonators B_(SH) and the parallel resonance frequency(f_(p,SER)) of the series resonators. The ladder network 40 providesgood attenuation below the series resonance frequency (f_(s,SH)) of theshunt resonators B_(SH) and above the parallel resonance frequency(f_(p,SER)) of the series resonators B_(SER). As noted previously, thereis a constant need to improve the performance of acoustic resonators andfilters that are based thereon.

Those skilled in the art will appreciate the scope of the presentdisclosure and realize additional aspects thereof after reading thefollowing detailed description of the preferred embodiments inassociation with the accompanying drawing figures.

SUMMARY

The present disclosure relates to tunable filter circuitry including aseries acoustic resonator between first and second nodes and acompensation circuit in parallel with the series acoustic resonator. Thecompensation circuit includes first and second inductors coupled inseries between the first node and the second node, wherein the firstinductor and the second inductor are negatively coupled with one anotherand a common node is provided between the first and second inductors.The compensation circuit also includes first and second shunt acousticresonators, which are coupled in parallel with one another between thecommon node and a fixed voltage node. A first variable capacitor is alsocoupled between the common node and the fixed voltage node, whereinchanging a capacitance of the first variable capacitor changes abandwidth of a passband of the filter circuitry. A first switch may becoupled in series with the second shunt acoustic resonator, wherein thefirst switch and the second shunt acoustic resonator are coupled betweenthe common node and the fixed voltage node.

The compensation circuit may also include a second variable capacitor,which is coupled in parallel with the at least one series acousticresonator, wherein changing a capacitance of the second variablecapacitor further changes the bandwidth of a passband of the filtercircuitry. Bias circuitry may be included to provide a DC bias to one ormore of the first and second shunt acoustic resonators, the seriesacoustic resonators, the first variable capacitor, and the secondvariable capacitor.

In one embodiment, a main series resonance is provided between the firstnode and the second node at a main resonance frequency through theseries acoustic resonator. First and second series resonances at firstand second resonance frequencies are provided between the first node andthe second node through the compensation circuit, wherein the first andsecond resonance frequencies are different.

BRIEF DESCRIPTION OF THE DRAWING FIGURES

The accompanying drawing figures incorporated in and forming a part ofthis specification illustrate several aspects of the disclosure and,together with the description, serve to explain the principles of thedisclosure.

FIG. 1 illustrates a conventional Bulk Acoustic Wave (BAW) resonator.

FIG. 2 is a graph of the magnitude and phase of impedance over frequencyresponses as a function of frequency for an ideal BAW resonator.

FIGS. 3A-3C are graphs of phase responses for various BAW resonatorconfigurations.

FIG. 4 illustrates a conventional BAW resonator with a border ring.

FIG. 5A is a schematic of a conventional ladder network.

FIGS. 5B and 5C are graphs of a frequency response for BAW resonators inthe conventional ladder network of FIG. 5A and a frequency response forthe conventional ladder network of FIG. 5A.

FIGS. 6A-6E are circuit equivalents for the ladder network of FIG. 5A atthe frequency points 1, 2, 3, 4, and 5, which are identified in FIG. 5C.

FIG. 7 illustrates an acoustic resonator in parallel with a compensationcircuit, which includes a single shunt acoustic resonator.

FIG. 8 is a graph that illustrates exemplary frequency responses for theacoustic resonator, compensation circuit, and overall filter circuit ofFIG. 7.

FIG. 9 illustrates an acoustic resonator in parallel with a compensationcircuit, which includes at least two shunt acoustic resonators,according to a first embodiment.

FIG. 10 is a graph that illustrates exemplary frequency responses forthe acoustic resonator, compensation circuit, and overall filter circuitof FIG. 9.

FIG. 11 is a graph that compares actual frequency responses of thefilter circuits of FIGS. 7 and 9.

FIG. 12 illustrates a plurality of parallel acoustic resonators inparallel with a compensation circuit, which includes at least two shuntacoustic resonators, according to a second embodiment.

FIG. 13 is a graph that illustrates first exemplary frequency responsesfor the acoustic resonator, compensation circuit, and overall filtercircuit of FIG. 12.

FIG. 14 is a graph that illustrates second exemplary frequency responsesfor the acoustic resonator, compensation circuit, and overall filtercircuit of FIG. 12.

FIGS. 15A through 15D illustrate transformation of the T-circuitimpedance architecture of the compensation circuit of FIG. 9 to a π (pi)impedance model.

FIG. 16 illustrates the filter circuit of FIG. 9 using the π (pi)impedance model of FIG. 15D.

FIG. 17 is a graph illustrating the overall shunt impedance, Zres,according to one embodiment.

FIG. 18 is a graph illustrating the series equivalent impedance, ZA,according to one embodiment.

FIGS. 19A and 19B are graphs over different frequency rangesillustrating the absolute or magnitude of series impedance ZS, theseries equivalent impedance ZA, and overall series impedance ZAs,according to one embodiment.

FIG. 20 illustrates an acoustic resonator in parallel with acompensation circuit including at least two shunt acoustic resonators,according to a third embodiment.

FIG. 21 illustrates a series resonant inductor-capacitor (L-C) circuitin parallel with a compensation circuit including at least two shuntacoustic resonators, according to a fourth embodiment.

FIG. 22 illustrates an acoustic resonator in parallel with acompensation circuit including at least two shunt acoustic resonators,according to a fifth embodiment.

FIG. 23 illustrates an acoustic resonator in parallel with acompensation circuit including at least two shunt acoustic resonators,according to a sixth embodiment.

FIG. 24 illustrates an acoustic resonator in parallel with acompensation circuit including at least two shunt acoustic resonators,according to a seventh embodiment.

FIG. 25 illustrates two series acoustic resonators in parallel with acompensation circuit including at least two shunt acoustic resonators,according to an eighth embodiment.

FIG. 26 illustrates two series acoustic resonators in parallel with acompensation circuit including at least two shunt acoustic resonators,according to a ninth embodiment.

FIG. 27 illustrates a communication circuit that is configured toprovide a tunable passband for the filter circuit, according to a tenthembodiment.

FIG. 28 is a graph illustrating a variable frequency response and theassociated return loss for the embodiment of FIG. 27.

FIG. 29 illustrates a communication circuit that is configured toprovide a tunable passband for the filter circuit, according to aneleventh embodiment.

FIG. 30 illustrates a communication circuit that is configured toprovide a tunable passband for the filter circuit, according to atwelfth embodiment.

FIGS. 31 and 32 are graphs illustrating a variable frequency responseand the associated return loss for a first example of the embodiment ofFIG. 30.

FIGS. 33 and 34 are graphs illustrating a variable frequency responseand the associated return loss for a second example of the embodiment ofFIG. 30.

FIG. 35 illustrates a communication circuit that is configured toprovide a tunable passband for the filter circuit, according to athirteenth embodiment.

FIG. 36 illustrates a communication circuit that is configured toprovide a tunable passband for the filter circuit, according to afourteenth embodiment.

DETAILED DESCRIPTION

The embodiments set forth below represent the necessary information toenable those skilled in the art to practice the embodiments andillustrate the best mode of practicing the embodiments. Upon reading thefollowing description in light of the accompanying drawing figures,those skilled in the art will understand the concepts of the disclosureand will recognize applications of these concepts not particularlyaddressed herein. It should be understood that these concepts andapplications fall within the scope of the disclosure and theaccompanying claims.

It will be understood that, although the terms first, second, etc. maybe used herein to describe various elements, these elements should notbe limited by these terms. These terms are only used to distinguish oneelement from another. For example, a first element could be termed asecond element, and similarly, a second element could be termed a firstelement, without departing from the scope of the present disclosure. Asused herein, the term “and/or” includes any and all combinations of oneor more of the associated listed items.

It will be understood that when an element such as a layer, region, orsubstrate is referred to as being “on” or extending “onto” anotherelement, it can be directly on or extend directly onto the other elementor intervening elements may also be present. In contrast, when anelement is referred to as being “directly on” or extending “directlyonto” another element, there are no intervening elements present.Likewise, it will be understood that when an element such as a layer,region, or substrate is referred to as being “over” or extending “over”another element, it can be directly over or extend directly over theother element or intervening elements may also be present. In contrast,when an element is referred to as being “directly over” or extending“directly over” another element, there are no intervening elementspresent. It will also be understood that when an element is referred toas being “connected” or “coupled” to another element, it can be directlyconnected or coupled to the other element or intervening elements may bepresent. In contrast, when an element is referred to as being “directlyconnected” or “directly coupled” to another element, there are nointervening elements present.

Relative terms such as “below” or “above” or “upper” or “lower” or“horizontal” or “vertical” may be used herein to describe a relationshipof one element, layer, or region to another element, layer, or region asillustrated in the figures. It will be understood that these terms andthose discussed previously are intended to encompass differentorientations of the device in addition to the orientation depicted inthe figures.

The terminology used herein is for the purpose of describing particularembodiments only and is not intended to be limiting of the disclosure.As used herein, the singular forms “a,” “an,” and “the” are intended toinclude the plural forms as well, unless the context clearly indicatesotherwise. It will be further understood that the terms “comprises,”“comprising,” “includes,” and/or “including” when used herein specifythe presence of stated features, integers, steps, operations, elements,and/or components but do not preclude the presence or addition of one ormore other features, integers, steps, operations, elements, components,and/or groups thereof.

Unless otherwise defined, all terms (including technical and scientificterms) used herein have the same meaning as commonly understood by oneof ordinary skill in the art to which this disclosure belongs. It willbe further understood that terms used herein should be interpreted ashaving a meaning that is consistent with their meaning in the context ofthis specification and the relevant art and will not be interpreted inan idealized or overly formal sense unless expressly so defined herein.

The present disclosure relates to tunable filter circuitry including aseries acoustic resonator between first and second nodes and acompensation circuit in parallel with the series acoustic resonator. Thecompensation circuit includes first and second inductors coupled inseries between the first node and the second node, wherein the firstinductor and the second inductor are negatively coupled with one anotherand a common node is provided between the first and second inductors.The compensation circuit also includes first and second shunt acousticresonators, which are coupled in parallel with one another between thecommon node and a fixed voltage node. A first variable capacitor is alsocoupled between the common node and the fixed voltage node, whereinchanging a capacitance of the first variable capacitor changes abandwidth of a passband of the filter circuitry. A first switch may becoupled in series with the second shunt acoustic resonator, wherein thefirst switch and the second shunt acoustic resonator are coupled betweenthe common node and the fixed voltage node.

The compensation circuit may also include a second variable capacitor,which is coupled in parallel with the at least one series acousticresonator wherein changing a capacitance of the second variablecapacitor further changes the bandwidth of a passband of the filtercircuitry. Bias circuitry may be provided to provide a DC bias to one ormore of the first and second shunt acoustic resonators, the seriesacoustic resonators, the first variable capacitor, and the secondvariable capacitor.

In various embodiments, the compensation circuit provides three primaryfunctions. The first is to provide a negative capacitive behavior, suchthat a negative capacitance is presented in parallel with the at leastone series acoustic resonator. As such, the effective capacitance of theat least one series acoustic resonator is reduced, which functions toshift the parallel resonance frequency f_(p) higher. The second functionis to add one or more additional series resonances between the first andsecond nodes. The combination of shifting the parallel resonancefrequency f_(p) higher and adding additional series resonances throughthe compensation circuit allows for passbands of greater bandwidth whilemaintaining excellent out-of-band rejection. The third function is todynamically control the bandwidth of the passbands. Details are providedbelow.

Turning now to FIG. 7, a series resonator B1 is shown coupled between aninput node I/P and an output node O/P. The series resonator B1 has aseries resonance frequency f_(s) and inherent capacitance, whichgenerally limits the bandwidth of filters that employ the seriesresonator B1. In the case of a Bulk Acoustic Wave (BAW) resonator, thecapacitance of the series resonator B1 is primarily caused by itsinherent structure, which looks and acts like a capacitor, in partbecause the series resonator includes the top and bottom electrodes 20,22 (FIG. 1) that are separated by a dielectric piezoelectric layer 18.While BAW resonators are the focus of the example, other types ofacoustic resonators, such as Surface Acoustic Wave (SAW) resonators, areequally applicable.

A compensation circuit 42 is coupled in parallel with the seriesresonator B1 and functions to compensate for some of the capacitancepresented by the series resonator B1. The compensation circuit 42includes two negatively coupled inductors L1, L2 and a shunt resonatorB2. The inductors L1, L2 are coupled in series between the input node VPand the output node O/P, wherein a common node CN is provided betweenthe inductors L1, L2. The inductors L1, L2 are magnetically coupled by acoupling factor K, wherein the dots illustrated in association with theinductors L1, L2 indicate that the magnetic coupling is negative. Assuch, the inductors L1, L2 are connected in electrical series andnegatively coupled from a magnetic coupling perspective. As definedherein, two (or more) series-connected inductors that are negativelycoupled from a magnetic perspective are inductors that are:

-   -   connected in electrical series; and    -   the mutual inductance between the two inductors functions to        decrease the total inductance of the two (or more) inductors.        The shunt resonator B2 is coupled between the common node CN and        ground, or other fixed voltage node.

To compensate for at least some of the capacitance of the seriesresonator B1, the compensation circuit 42 presents itself as a negativecapacitance within certain frequency ranges, when coupled in parallelwith the series resonator B1. Since capacitances in parallel areadditive, providing a negative capacitance in parallel with the(positive) capacitance of the series resonator B1 effectively reducesthe capacitance of the series resonator B1. With the compensationcircuit 42, the series resonator B1 can actually function as a filter(instead of just a resonator) and provide a passband, albeit a fairlynarrow passband, instead of a more traditional resonator response (solidline of FIG. 2). FIG. 8 graphically illustrates the frequency responsesof the series resonator B1 (inside the block referenced B1), thecompensation circuit 42 (inside the block referenced 42), and the filtercircuit in which the compensation circuit 42 is placed in parallel withthe series resonator B1. As illustrated, the filter circuit provides arelatively narrow passband. Further detail on this particular circuittopology can be found in the co-assigned U.S. patent application Ser.No. 15/004,084, filed Jan. 22, 2016, and titled RF LADDER FILTER WITHSIMPLIFIED ACOUSTIC RF RESONATOR PARALLEL CAPACITANCE COMPENSATION, andU.S. patent application Ser. No. 14/757,651, filed Dec. 23, 2015, andtitled SIMPLIFIED ACOUSTIC RF RESONATOR PARALLEL CAPACITANCECOMPENSATION, which are incorporated herein by reference.

While beneficial in many applications, the narrow passband of thecircuit topology of FIG. 7 has its limitations. With the challenges ofmodern day communication systems, wider passbands and the ability toprovide multiple passbands within a given system are needed.Fortunately, Applicants have discovered that certain modifications tothis topology provide significant and truly unexpected increases inpassband bandwidths and, in certain instances, the ability to generatemultiple passbands of the same or varying bandwidths in an efficient andeffective manner.

With reference to FIG. 9, a modified circuit topology is illustratedwherein the circuit topology of FIG. 7 is modified to include anadditional shunt resonator B3, which is coupled between the common nodeCN and ground. As such, a new compensation circuit 44 is created thatincludes the negatively coupled inductors L1 and L2, which have acoupling coefficient K, and at least two shunt resonators B2, B3. Thecompensation circuit 44 is coupled in parallel with the series resonatorB1. When the series resonance frequencies f_(s) of the shunt resonatorsB2, B3 are different from one another, unexpectedly wide bandwidthspassbands are achievable while maintaining a very flat passbands, steepskirts, and excellent cancellation of signals outside of the passbands.

FIG. 10 graphically illustrates the frequency responses of the seriesresonator B1 (inside the block referenced B1), the compensation circuit44 (inside the block referenced 44), and the filter circuit in which thecompensation circuit 44 is placed in parallel with the series resonatorB1. As illustrated, the filter circuit with compensation circuit 44provides a much wider passband (FIG. 10) than the filter circuit withcompensation circuit 42 (FIG. 8).

While FIGS. 8 and 10 are graphical representations, FIG. 11 is an actualcomparison of the frequency response of the filter circuit using thedifferent compensation circuits 42, 44, wherein the filter circuit usingcompensation circuit 44 provides a significantly wider and better formedpassband (solid line) than the filter circuit using compensation circuit42 (dashed line).

As illustrated in FIG. 12, the concepts described herein not onlycontemplate the use of multiple shunt resonators B2, B3, which arecoupled between the common node CN and ground, but also multiple seriesresonators, such as series resonators B1 and B4, which are coupled inparallel with one another between the input node VP and the output nodeO/P. The series resonance frequencies f_(s) of the series resonators B1,B4 are different from one another, and the series resonance frequenciesf_(s) of the shunt resonators B2, B3 are also different from one anotherand different from those of the series resonators. While only two seriesresonators B1, B4 and two shunt resonators B2, B3 are illustrated, anynumber of these resonators may be employed depending on the applicationand the desired characteristics of the overall frequency response of thecircuit in which these resonators and associated compensation circuits44 are employed. While the theory of operation is described furthersubsequently, FIGS. 13 and 14 illustrate just two of the manypossibilities.

For FIG. 13, there are two series resonators B1, B4 and two shuntresonators B2, B3, with different and relatively dispersed seriesresonance frequencies f_(s). FIG. 13 graphically illustrates thefrequency responses of the combination of the two series resonators B1,B4 (inside the block referenced BX), the compensation circuit 44 withtwo shunt resonators B2, B3 (inside the block referenced 44), and thefilter circuit in which the compensation circuit 44 is placed inparallel with the series resonators B1, B4. As illustrated, the filtercircuit in this configuration has the potential to provide a passbandthat is even wider than that for the embodiment of FIGS. 9 and 10. Forexample, passbands of greater than 100 MHz, 150 MHz, 175 MHz, and 200MHz are contemplated at frequencies at or above 1.5 GHz, 1.75 GHz, and 2GHz, respectively. In other words, center-frequency-to-bandwidth ratios(fc/BW*100) of 3.5% to 9%, 12%, or greater are possible, wherein fc isthe center frequency of the passband and BW is the bandwidth of thepassband. If multiple passbands are provided, BW may encompass all ofthe provided passbands. Further, when multiple passbands are provided,the passbands may have the same or different bandwidths orcenter-frequency-to-bandwidth ratios. For example, one passband may havea relatively large center-frequency-to-bandwidth ratio, such as 12%, anda second passband may have a relatively smallcenter-frequency-to-bandwidth ratio, such as 2%. Alternatively, multipleones of the passbands may have a bandwidth of 100 MHz, or multiple onesof the passbands may have generally the samecenter-frequency-to-bandwidth ratios. In the latter case, the bandwidthsof the passbands may inherently be different from one another, eventhough the center-frequency-to-bandwidth ratios are the same.

For FIG. 14, there are four series resonators, which are coupled inparallel with one another (not shown), and two shunt resonators (notshown) with different and more widely dispersed series resonancefrequencies f_(s). FIG. 14 graphically illustrates the frequencyresponses of the combination of the four series resonators (inside theblock referenced BX), the compensation circuit 44 with two shuntresonators B2, B3 (inside the block referenced 44), and the filtercircuit in which the compensation circuit 44 is placed in parallel withfour series resonators. As illustrated, the filter circuit in thisconfiguration provides multiple passbands, which are separated by a stopband. In this embodiment, two passbands are provided; however, thenumber of passbands may exceed two. The number of passbands and thebandwidth of each of the passbands is a function of the number of shuntand series resonators B1-B4 and the series resonance frequencies f_(s)thereof.

The theory of the compensation circuit 44 follows and is described inassociation with FIGS. 15A through 15D and 16. With reference to FIG.15A, assume the compensation circuit 44 includes the two negativelycoupled inductors L1, L2, which have an inductance value L, and two ormore shunt resonators BY, which have an overall shunt impedance Zrespresented between the common node CN and ground. While the inductancevalues L of the negatively coupled inductors L1, L2 are described asbeing the same, these values may differ depending on the application.Also assume that the one or more series resonators BX present an overallseries impedance ZS.

As shown in FIG. 15B, the two negatively coupled and series-connectedinductors L1, L2 (without Zres) can be modeled as a T-network of threeinductors L3, L4, and L5, wherein series inductors L3 and L4 areconnected in series and have a value of L(1+K), and shunt inductor L5has a value of −L*K, where K is a coupling factor between the negativelycoupled inductors L1, L2. Notably, the coupling factor K is a positivenumber between 0 and 1. Based on this model, the overall impedance ofthe compensation circuit 44 is modeled as illustrated in FIG. 15C,wherein the shunt impedance Zres is coupled between the shunt inductorL5 and ground. The resulting T-network, as illustrated in FIG. 15C, canbe transformed into an equivalent π (pi) network, as illustrated in FIG.15D.

The π network of FIG. 15D can be broken into a series impedance ZA andtwo shunt equivalent impedances ZB. The series equivalent impedance ZAis represented by two series inductances of value L*(1+K), where K>0,and a special “inversion” impedance Zinv. The inversion impedance Zinvis equal to [L(1+K)ω]²/[Zres−jLKω], where ω=2πf and f is the frequency.As such, the series equivalent impedance ZA equals j*2*L(1+K)ω+Zinv andis coupled between the input node I/O and the output node O/P. Each ofthe two shunt equivalent impedances ZB is represented by an inductor ofvalue L(1−K) in series with two overall shunt impedances Zres.

Notably, the series equivalent impedance ZA has a negative capacitorbehavior at certain frequencies at which broadband cancellation isdesired and has series resonance at multiple frequencies. In general,the series equivalent impedance ZA has a multiple bandpass-bandstopcharacteristic in that the series equivalent impedance ZA will pass somefrequencies and stop others. When the series equivalent impedance ZA isplaced in parallel with the series impedance ZS of the series resonatorsBX, which can also have a multiple bandpass-bandstop characteristic, abroadband filter or a filter with multiple passbands may be created.

FIG. 16 illustrates the series impedance ZS of the series resonators BXin parallel with the series equivalent impedance ZA of the compensationcircuit 44. The overall series impedance ZAs represents the seriesimpedance ZS in parallel with the series equivalent impedance ZA. Thetwo shunt impedances ZB are respectively coupled between the input portVP and ground and the output port O/P and ground. The primary focus forthe following discussion relates to the series equivalent impedance ZAand its impact on the series impedance ZS when the series equivalentimpedance ZA is placed in parallel with the series impedance ZS.

As noted previously, the series equivalent impedance ZA provides twoprimary functions. The first provides a negative capacitive behavior,and the second provides one or more additional series resonances betweenthe input node VP and the output node O/P. These additional seriesresonances are provided through the series equivalent impedance ZA andare in addition to any series resonances that are provided through theseries impedance ZS of the series resonators BX. To help explain thebenefits and concept of the negative capacitive behavior provided by theseries equivalent impedance ZA, normal capacitive behavior isillustrated in association with the overall shunt impedance Zres, whichis provided by the shunt resonators BY. FIG. 17 graphs the absolute(magnitude) and imaginary components of the overall shunt impedanceZres, which is formed by two shunt resonators BY, which are coupled inparallel with one another.

The series resonance frequency f_(s) for each of the two shuntresonators BY occurs when the absolute impedance (abs(Zres)) is at ornear zero. Since there are two shunt resonators BY, the absoluteimpedance (abs(Zres)) is at or near zero at two frequencies, and assuch, there are two series resonance frequencies f_(s). The parallelresonance frequencies f_(p) occur when the imaginary component(imag(Zres)) peaks. Again, since there are two shunt resonators BY,there are two parallel resonance frequencies f_(p) provided by theoverall shunt impedance Zres.

Whenever the imaginary component (imag(Zres)) of the overall shuntimpedance Zres is less than zero, the overall shunt impedance Zres has acapacitive behavior. The capacitive behavior is characterized in thatthe reactance of the overall shunt impedance Zres is negative anddecreases as frequency increases, which is consistent with capacitivereactance, which is represented by 1/jωC. The graph of FIG. 17identifies three regions within the impedance response of the overallshunt impedance Zres that exhibit capacitive behavior.

Turning now to FIG. 18, the series equivalent impedance ZA isillustrated over the same frequency range as that of the overall shuntimpedance Zres, which was illustrated in FIG. 17. The series equivalentimpedance ZA has two series resonance frequencies f_(s), which occurwhen the absolute impedance (abs(ZA)) is at or near zero. The two seriesresonance frequencies f_(s) for the series equivalent impedance ZA aredifferent from each other and slightly different from those for theoverall shunt impedance Zres. Further, the number of series resonancefrequencies f_(s) generally corresponds to the number of shuntresonators BY in the compensation circuit 44, assuming the seriesresonance frequencies f_(s) are different from one another.

Interestingly, the imaginary component (imag(ZA)) of the seriesequivalent impedance ZA is somewhat inverted with respect to that of theoverall shunt impedance Zres. Further, the imaginary component(imag(ZA)) of the series equivalent impedance ZA has a predominantlypositive reactance. During the portions at which the imaginary component(imag(ZA)) is positive, the reactance of the series equivalent impedanceZA again decreases as frequency increases, which is indicative ofcapacitive behavior. However, the reactance is positive, whereastraditional capacitive behavior would present a negative reactance. Thisphenomenon is referred to as negative capacitive behavior. Thoseportions of the imaginary component (imag(ZA)) of the series equivalentimpedance ZA that are positive and thus exhibit negative capacitivebehavior are highlighted in the graph of FIG. 18.

The negative capacitive behavior of the series equivalent impedance ZAfor the compensation circuit 44 is important, because when the seriesequivalent impedance ZA is placed in parallel with the series impedanceZS, the effective capacitance of the filter circuit is reduced. Reducingthe effective capacitance of the filter circuit shifts the parallelresonance frequency f_(p) of the series impedance ZS higher in thefrequency range, which is described subsequently, and significantlyincreases the available bandwidth for passbands while providingexcellent out-of-band rejection.

An example of the benefit is illustrated in FIGS. 19A and 19B. The solidline, which is labeled abs(VG), represents the frequency response of thefilter circuit illustrated in FIG. 12, wherein there are two seriesresonators BX and two shunt resonators BY in the compensation circuit44. The frequency response has two well-defined passbands, which areseparated by a stop band. The frequency response abs(VG) of the filtercircuit generally corresponds to the inverse of the overall seriesimpedance ZAs, which again represents the series impedance ZS inparallel with the series equivalent impedance ZA, as provided in FIG.16.

Notably, the parallel resonance frequencies f_(p)(ZS) of the seriesimpedance ZS, in isolation, fall in the middle of the passbands offrequency response abs(VG) of the filter circuit. If the parallelresonance frequencies f_(p)(ZS) of the series impedance ZS remained atthese locations, the passbands would be severely affected. However, thenegative capacitive behavior of the series equivalent impedance ZAfunctions to shift these parallel resonance frequencies f_(p)(ZS) of theseries impedance ZS to a higher frequency and, in this instance, abovethe respective passbands. This is manifested in the resulting overallseries impedance ZAs, in which the only parallel resonance frequenciesf_(p)(ZAs) occur above and outside of the respective passbands. Anadditional benefit to having the parallel resonance frequenciesf_(p)(ZAs) occur outside of the respective passbands is the additionalcancellation of frequencies outside of the passbands. Plus, the overallseries impedance ZAs is lower than the series impedance ZS within therespective passbands.

A further contributor to the exemplary frequency response abs(VG) of thefilter circuit is the presence of the additional series resonancefrequencies f_(s), which are provided through the series equivalentimpedance ZA. These series resonance frequencies f_(s) are offset fromeach other and from those provided through the series impedance ZS. Theseries resonance frequencies f_(s) for the series equivalent impedanceZA in the series impedance ZS occur when the magnitudes of therespective impedances approach zero. The practical results are widerpassbands, steeper skirts for the passbands, and greater rejectionoutside of the passbands, as evidenced by the frequency response abs(VG)of the filter circuit.

Turning now to FIG. 20, another embodiment is provided wherein thecompensation circuit 44 is placed in parallel with one or more seriesresonators BX. In this embodiment, shunt resonator B2 is permanentlycoupled between the common node CN and ground. Shunt resonators B3, B5,and B6 can be selectively coupled between the common node CN and groundvia respective switches S1, S2, and S3. By using control circuitry (notshown) to selectively switch the various shunt resonators B3, B5, and B6into and out of the compensation circuit 44, the passbands and stopbands provided by the filter circuit can be dynamically adjusted fordifferent modes of operation. Again, the series resonance frequenciesf_(s) of the shunt resonators B2, B3, B5, and B6 will generally differfrom one another. Resistors R1 and R2 are illustrated and may be coupledbetween the input node I/P and ground and the output node O/P andground, respectively. In one embodiment, the series resonance frequencyf_(s) of the series equivalent impedance ZA is greater than the seriesresonance frequency f_(s) of the series impedance ZS. The seriesresonance frequency f_(s) of at least one of the shunt resonators B2,B3, B5, and B6 is greater than the series resonance frequency f_(s) ofthe series impedance ZS. As with any of these embodiments, the number ofshunt resonators BY and series resonators BX may vary from embodiment toembodiment. The number illustrated is merely for illustrative purposes.

FIG. 21 illustrates yet another embodiment, which is similar to thatillustrated in FIG. 20. The difference is that the series resonators BXare replaced with a lumped series L-C circuit, which is formed from aseries capacitor CS and a series inductor LS that are coupled in seriesbetween the input node I/P and the output node O/P.

FIG. 22 illustrates an embodiment similar to that of FIG. 9, except thatat least one inductor L6 is coupled in series with shunt resonator B3.As such, shunt resonator B2 is coupled between the common node CN andground without a series inductor, and shunt resonator B2 and inductor L6are coupled in series with one another and between the common node CNand ground. In one embodiment, the series resonance frequency f_(s) ofthe series equivalent impedance ZA is greater than the lowest seriesresonance frequency f_(s) of the series impedance ZS. FIG. 23illustrates a further modification to the embodiment of FIG. 22, whereinan inductor L7 is placed in series with the series resonators BX, suchthat the inductor L7 and the series resonators BX are coupled in serieswith one another between the input node I/P and the output node O/P.

With reference to FIG. 24, a more complex filter arrangement isillustrated. In particular, resonators B7, B8, and B9 are coupled inseries between the input node I/P and the output node O/P. Thecompensation circuits 44 of FIG. 22 are coupled across resonator B7 andresonator B9. The resonators B7, B8, and B9 may each represent a singleacoustic resonator or multiple acoustic resonators in parallel. Notably,the shunt resonators B2, B3, B5, B6 in the embodiments of FIGS. 21through 24 and 30 are considered to be in parallel with one anotherbetween the common node CN and ground, even if an additional switch,inductive, capacitive, or like element is provided in series with one ormore of the shunt resonators.

FIG. 25 illustrates yet another embodiment wherein resonators B10 andB11 are coupled in series between the input node I/P and the output nodeO/P. An inductor L8 is coupled between node N1 and ground. Thecompensation circuit 44 is coupled across both of the resonators B10 andB11. Accordingly, the compensation circuit 44 may be coupled across oneor more acoustic resonators along the series path that extends betweenthe input node I/P and the output node O/P.

FIG. 26 illustrates a modification to the embodiment of FIG. 25, whereinthe inductor L8 is replaced with a shunt resonator B12.

FIG. 27 illustrates an embodiment wherein the compensation circuit 44 istunable, such that the bandwidth of the passband for the filter circuitcan be varied, or tuned, in a dynamic fashion. As illustrated, thecompensation circuit 44 is provided in parallel with the seriesresonator BX. The compensation circuit 44 differs from theabove-described embodiments in that a variable capacitor C1, orvaractor, is placed in parallel with at least two of the shuntresonators B2, B3 between the common node CN and ground. As thecapacitance of the variable capacitor C1 varies, the bandwidth of thepassband for the filter circuit will vary. In this embodiment, the upperskirt of the passband will remain relatively constant, while thelocation of the lower skirt will vary. In particular, as the capacitanceof the variable capacitor C1 increases, the location of the lower skirtof the passband increases, and vice versa.

The graph of FIG. 28 illustrates the frequency responses of the filtercircuit of FIG. 27 at three different capacitance levels for thevariable capacitor C1. Frequency response FR1 has the broadest passbandbandwidth and corresponds to the lower of the three capacitance levels.Frequency response FR2 provides an intermediate passband bandwidth andcorresponds to an intermediate capacitance level for the variablecapacitor C1. Frequency response FR3 provides the narrowest passbandbandwidth and corresponds to a higher capacitance level for the variablecapacitor C1. As depicted, the upper skirts of the three passbandsremain relatively constant, and the location of the lower skirts for thethree passbands progressively increase as the capacitance provided bythe variable capacitor C1 increases. The result is that the bandwidth ofthe passbands decrease as the capacitance provided by the variablecapacitor C1 increases.

As illustrated in FIG. 28, the three frequency responses FR1-3 exhibitflat passbands, steep side skirts, and excellent out-of-band rejectionabove and below the passbands. A further benefit of the describedcircuitry is that the return losses of RL1-3 are excellent within thepassband for each of the three corresponding frequency responses FR1-3,especially for the two wider-bandwidth passbands (RL1 and RL2).

FIG. 29 illustrates a modified version of the filter circuit of FIG. 28.In this example, an additional variable capacitor C2 is provided inparallel with at least two series resonators B1, B4. The additionalvariable capacitor C2 provides additional tunability of the overallfrequency response.

Turning now to FIG. 30, another embodiment is provided where theadditional variable capacitor C2 is provided in parallel with at leasttwo series resonators B1, B4. In this embodiment, shunt resonator B2 ispermanently coupled between the common node CN and ground. Shuntresonators B3 and B5 can be selectively coupled between the common nodeCN and ground via respective switches S1 and S2. Variable capacitor C1is provided in parallel with the shunt resonators B3 and B5 between thecommon node CN and ground. By using control circuitry 46 to selectivelyswitch the various shunt resonators B3 and B5 into and out of thecompensation circuit 44 using switches S1 and S2 as well as varying thecapacitance of the variable capacitors C1 and C2, the bandwidths andlocations of the passbands provided by the filter circuit can bedynamically adjusted for different modes of operation. Again, the seriesresonance frequencies f_(s) of the shunt resonators B2, B3 and B5 willgenerally differ from one another. In one embodiment, the seriesresonance frequency f_(s) of at least one of the shunt resonators B2,B3, and B5 is greater than the series resonance frequency f_(s) of theseries impedance ZS. As with any of these embodiments, the number ofshunt resonators BY and series resonators BX may vary from embodiment toembodiment. The number illustrated is merely for illustrative purposes.

FIGS. 31 and 32 are graphs of frequency responses and return losses foran exemplary configuration of the filter circuit of FIG. 30. Assume thatthe shunt resonator B2 has a series resonance frequency of 2.700 GHz,shunt resonator B3 has a series resonance frequency of 2.470 GHz, andshunt resonator B5 as a series resonance frequency of 2.490 GHz. For thefilter circuit of FIG. 30, shunt resonator B2 is permanently coupledbetween the common node CN and ground, and the shunt resonators B3 andB5 are alternately switched into and out of the circuit using switchesS1 and S2, respectively. The total inherent, or parasitic, capacitancefor either of the shunt resonators B3 or B5 in parallel with shuntresonator B2 is approximately 0.65 pF. Assume that the variablecapacitor C1 has a capacitance that can vary between 0 pF and at least0.20 pF.

The frequency response FR4 and the return loss RL4 illustrated in FIG.31 corresponds to the shunt resonator B3 being switched into thecircuit, shunt resonator B5 being switched out of the circuit, and thecapacitance for the variable capacitor C1 being set to zero. Thepassband provided by the frequency response FR4 corresponds to therequisite passband for downlink band 41 (B41) of the LTE (Long TermEvolution) standard for cellular communications. The passband fordownlink LTE band B41 is 2.496 GHz to 2.690 GHz.

The frequency response can be dynamically modified to the requisitepassband for downlink band 38X (B38X) of the LTE standard by switchingthe shunt resonator B3 out of the circuit, switching the shunt resonatorB5 into the circuit, and adjusting the capacitance for the variablecapacitor C1 to approximately 0.20 pF, such that the total capacitancebetween the common node CN and ground is approximately 0.85 pF (0.20pF+0.65 pF). The passband for downlink LTE band B38X is 2.545 GHz to2.655 GHz, wherein the upper end of the passband is only 35 MHz lowerthat required for LTE band B41. In FIG. 32, the frequency response FR5and the return loss RL5 for LTE band B38X is illustrated along with thefrequency response FR4 and the return loss RL4 for LTE band B41. Asillustrated, LTE bands B41 and B38X have similar upper skirts but havesignificantly different locations for the lower skirts. Notably, thereturn losses RL4 and RL5 within the passbands are exceptionally low,and in this example, less than −13 dB throughout a vast majority of thepassbands. In this example, assume that variable capacitor C2 is notused or is set to approximately 0 pF.

For the example associated with graphs of FIGS. 33 and 34, the variablecapacitor C2 is employed along with the variable capacitor C1. Assumethat the shunt resonator B2 has a series resonance frequency of 2.700GHz, shunt resonator B3 has a series resonance frequency of 2.470 GHz,shunt resonator B5 as a series resonance frequency of 2.550 GHz, andseries resonator B1 as a series resonance frequency of 2.552 GHz. Again,shunt resonator B2 is permanently coupled between the common node CN andground, and the shunt resonators B3 and B5 are alternately switched intoand out of the circuit using switches S1 and S2, respectively. The totalinherent, or parasitic, capacitance for either of the shunt resonatorsB3 or B5 in parallel with shunt resonator B2 is approximately 0.95 pF.Assume that the variable capacitor C1 has a capacitance that can varybetween 0 pF and at least 1.20 pF, and the variable capacitor C1 has acapacitance that can vary between 0 pF and at least 0.35 pF.

The frequency response FR6 and the return loss RL6 illustrated in FIG.33 correspond to the shunt resonator B3 being switched into the circuit,shunt resonator B5 being switched out of the circuit, and thecapacitances for the variable capacitor C1 and the variable capacitor C2being set to zero. The passband provided by the frequency response FR4again corresponds to the requisite passband for LTE band B41, which is2.496 GHz to 2.690 GHz.

The frequency response can be dynamically modified to the requisitepassband for downlink LTE band 7 (B7RX) of the LTE standard by switchingthe shunt resonator B3 out of the circuit, switching the shunt resonatorB5 into the circuit, and adjusting the capacitance for the variablecapacitor C1 to approximately 1.20 pF and the capacitance for thevariable capacitor C2 to approximately 0.35 pF. The total capacitancebetween the common node CN and ground is approximately 1.85 pF (1.20pF+0.65 pF), and the capacitance across the series resonators B1 and B4is approximately 1.3 pF (0.95 pF+0.35 pF). The passband for LTE bandB7RX is 2.620 GHz to 2.690 GHz, wherein the upper end of the passbandaligns precisely with that required for LTE band B41.

In FIG. 34, the frequency response FR7 and the return loss RL7 for LTEband B7RX is illustrated along with the frequency response FR6 and thereturn loss RL6 for LTE band B41. As illustrated, LTE bands B41 and B7RXhave upper skirts that align with one another, but have significantlydifferent locations for the lower skirts. Notably, the return losses RL6and RL7 within the passbands are exceptionally low, and this example,less than −13 dB throughout a vast majority of the passbands. While theabove examples relate to LTE bands B41, B41RX, and B38X, the conceptsabove can apply to any bands in any communication standard. The conceptsabove are particularly beneficial when implemented in receive filtersthat rest between one or more antennas and downconversion or otherreceiver circuitry in virtually any wireless communication application.These concepts may also be implemented in transmit filters.

With reference to FIG. 35, other techniques may be applied to thecompensation circuit 44 to facilitate tuning the passband for theassociated filter circuitry. As illustrated, bias circuitry 48 may beused to apply one or more DC bias signals to one or more of the variablecapacitor C1, shunt resonator B2, and shunt resonator B3. In thisexample, the DC bias signal is a DC voltage that is applied to thecommon node CN. Doing so adjusts the effective capacitance between thecommon node CN and ground, and thus, will shift at least the lower skirtof the passband up or down.

As illustrated in FIG. 36, the same DC bias signal or different DC biassignals may be provided to the variable capacitor C2, series resonatorB1, and/or series resonator B4, to adjust the series capacitancepresented between the input node VP and the output node O/P. Adjustingthe series capacitance also impacts the center frequency, lower skirt,and/or upper skirt of the passband provided by the filter circuit. Thebias circuitry 48 may also be applied to the embodiments of FIG. 30,wherein the DC bias signals provided to the variable capacitor C1,variable capacitor C2, and/or any of the series resonators B1, B4 andshunt resonators B2, B3, and B5.

Those skilled in the art will recognize numerous modifications and otherembodiments that incorporate the concepts described herein. Thesemodifications and embodiments are considered to be within scope of theteachings provided herein and the claims that follow.

What is claimed is:
 1. Filter circuitry comprising: a first node and asecond node; at least one series acoustic resonator coupled between thefirst node and the second node, wherein at least one main seriesresonance is provided between the first node and the second node at amain resonance frequency through the at least one series acousticresonator; and a compensation circuit comprising: a first inductor and asecond inductor coupled in series between the first node and the secondnode, wherein the first inductor and the second inductor are negativelycoupled with one another and a common node is provided between the firstinductor and the second inductor; a first shunt acoustic resonatorcoupled between the common node and a fixed voltage node; a second shuntacoustic resonator coupled between the common node and the fixed voltagenode, wherein a first series resonance at a first resonance frequencyand a second series resonance at a second resonance frequency, which isdifferent from the first resonance frequency and main resonancefrequency, are provided between the first node and the second nodethrough the compensation circuit; and a first variable capacitor coupledbetween the common node and the fixed voltage node, wherein changing acapacitance of the first variable capacitor changes a bandwidth of apassband of the filter circuitry.
 2. The filter circuitry of claim 1further comprising a second variable capacitor coupled in parallel withthe at least one series acoustic resonator, wherein changing acapacitance of the second variable capacitor changes the bandwidth ofthe passband of the filter circuitry.
 3. The filter circuitry of claim 2further comprising bias circuitry configured to apply at least one biassignal to at least one of the first shunt acoustic resonator, the secondshunt acoustic resonator, and the first variable capacitor, as well asto at least one of the second variable capacitor and the at least oneseries acoustic resonator, wherein changing the at least one bias signalchanges the bandwidth of the passband of the filter circuitry.
 4. Thefilter circuitry of claim 1 further comprising bias circuitry configuredto apply a bias signal to at least one of the first shunt acousticresonator, the second shunt acoustic resonator, and the first variablecapacitor, wherein changing the bias signal changes the bandwidth of thepassband of the filter circuitry.
 5. The filter circuitry of claim 1further comprising a first switch coupled in series with the secondshunt acoustic resonator, wherein the first switch and the second shuntacoustic resonator are coupled between the common node and the fixedvoltage node.
 6. The filter circuitry of claim 5 further comprising asecond variable capacitor coupled in parallel with the at least oneseries acoustic resonator, wherein changing a capacitance of the secondvariable capacitor changes the bandwidth of the passband of the filtercircuitry.
 7. The filter circuitry of claim 6 further comprising biascircuitry configured to apply at least one bias signal to at least oneof the first shunt acoustic resonator, the second shunt acousticresonator, and the first variable capacitor, as well as to at least oneof the second variable capacitor and the at least one series acousticresonator, wherein changing the at least one bias signal changes thebandwidth of the passband of the filter circuitry.
 8. The filtercircuitry of claim 1 wherein a frequency response of the filtercircuitry comprises a plurality of passbands such that adjacentpassbands of the plurality of passbands are separated by a stop band. 9.The filter circuitry of claim 1 wherein at least one of the firstresonance frequency and the second resonance frequency is greater thanthe main resonance frequency.
 10. The filter circuitry of claim 1wherein the first resonance frequency is less than the main resonancefrequency, and the second resonance frequency is greater than the mainresonance frequency.
 11. The filter circuitry of claim 1 wherein the atleast one series acoustic resonator comprises a plurality of acousticresonators that are coupled in parallel with one another, and each ofthe plurality of acoustic resonators has a different series resonancefrequency.
 12. The filter circuitry of claim 1 wherein the compensationcircuit comprises at least one additional shunt acoustic resonatorcoupled between the common node and the fixed voltage node.
 13. Thefilter circuitry of claim 1 wherein: an equivalent π (pi) network of thecompensation circuit comprises a series equivalent impedance between thefirst node and the second node as well as two shunt equivalentimpedances; and the series equivalent impedance exhibits negativecapacitive behavior throughout multiple frequency ranges.
 14. The filtercircuitry of claim 1 wherein: an equivalent π (pi) network of thecompensation circuit comprises a series equivalent impedance between thefirst node and the second node and two shunt equivalent impedances; andthe first series resonance at the first resonance frequency and thesecond series resonance at the second resonance frequency are providedthrough the series equivalent impedance.
 15. The filter circuitry ofclaim 1 wherein: an equivalent π (pi) network of the compensationcircuit comprises a series equivalent impedance between the first nodeand the second node and two shunt equivalent impedances; the at leastone series acoustic resonator comprises a series impedance having aparallel resonance at a first parallel resonance frequency; and theseries impedance of the at least one series acoustic resonator inparallel with the series equivalent impedance of the equivalent π (pi)network of the compensation circuit provides an overall impedance havinga parallel resonance at a second parallel resonance frequency, which isgreater than the first parallel resonance frequency.
 16. The filtercircuitry of claim 1 wherein: an equivalent π (pi) network of thecompensation circuit comprises a series equivalent impedance between thefirst node and the second node and two shunt equivalent impedances; theat least one series acoustic resonator comprises a series impedancehaving a parallel resonance at a first parallel resonance frequency; theseries impedance of the at least one series acoustic resonator inparallel with the series equivalent impedance of the equivalent π (pi)network of the compensation circuit provides an overall impedance havinga parallel resonance at a second parallel resonance frequency, which isgreater than the first parallel resonance frequency; and the seriesequivalent impedance exhibits negative capacitive behavior throughoutmultiple frequency ranges.
 17. The filter circuitry of claim 16 whereinat least one of the first resonance frequency and the second resonancefrequency is greater than the main resonance frequency.
 18. The filtercircuitry of claim 16 wherein the first resonance frequency is less thanthe main resonance frequency, and the second resonance frequency isgreater than the main resonance frequency.
 19. The filter circuitry ofclaim 1 wherein at least one of the first shunt acoustic resonator andthe second shunt acoustic resonator has a series resonance at a thirdresonance frequency wherein the third resonance frequency is greaterthan the main resonance frequency.
 20. The filter circuitry of claim 1wherein the first inductor and the second inductor have differentinductances.
 21. The filter circuitry of claim 1 wherein the at leastone series acoustic resonator, the first shunt acoustic resonator, andthe second shunt acoustic resonator are at least one of a bulk acousticwave (BAW) resonator and a surface acoustic wave (SAW) resonator. 22.The filter circuitry of claim 1 wherein fc/BW*100 is between 3.5% and12%, wherein fc is a center frequency of the passband of the filtercircuitry, and BW is the bandwidth of the passband.
 23. Filter circuitrycomprising: a first node and a second node; at least one series acousticresonator coupled between the first node and the second node, wherein atleast one main series resonance is provided between the first node andthe second node at a main resonance frequency through the at least oneseries acoustic resonator; and a compensation circuit comprising: afirst inductor and a second inductor coupled in series between the firstnode and the second node, wherein the first inductor and the secondinductor are negatively coupled with one another and a common node isprovided between the first inductor and the second inductor; a firstshunt acoustic resonator coupled between the common node and a fixedvoltage node; a second shunt acoustic resonator coupled between thecommon node and the fixed voltage node; and a first variable capacitorcoupled between the common node and the fixed voltage node, whereinchanging a capacitance of the first variable capacitor changes abandwidth of a passband of the filter circuitry.
 24. The filtercircuitry of claim 23 further comprising a second variable capacitorcoupled in parallel with the at least one series acoustic resonatorwherein changing a capacitance of the second variable capacitor changesthe bandwidth of the passband of the filter circuitry.